Pilot transmission and channel estimation for an OFDM system with excess delay spread

ABSTRACT

Pilot transmission and channel estimation techniques for an OFDM system with excess delay spread are described. To mitigate the deleterious effects of excess delay spread, the number of pilot subbands is greater than the cyclic prefix length. This “oversampling” may be achieved by using more pilot subbands in each symbol period or different sets of pilot subbands in different symbol periods. In one example, a first set of pilot subands may be received in a first symbol period, and a second set of pilot subands may be received in a second symbol period. The first set of pilot subands and the second set of pilot subbands may be staggered in frequency.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application for patent is a continuation of U.S. patentapplication Ser. No. 12/041,510 entitled “Pilot Transmission and ChannelEstimation for an OFDM System with Excess Delay Spread” filed Mar. 3,2008, now pending, which is a continuation of U.S. patent applicationSer. No. 10/821,706 entitled “Pilot Transmission and Channel Estimationfor an OFDM System with Excess Delay Spread” filed Apr. 9, 2004 now U.S.Pat. No. 7,339,999 granted Mar. 4, 2008 which claims priority toprovisional application No. 60/538,210 entitled “Pilot Transmission andChannel Estimation for an OFDM System with Excess Delay Spread” filedJan. 21, 2004, all assigned to the assignee hereof and hereby expresslyincorporated by reference herein in their entirely.

FIELD

The present invention relates generally to data communication, and morespecifically to pilot transmission and channel estimation for anorthogonal frequency division multiplexing (OFDM) system with excessdelay spread.

BACKGROUND

OFDM is a multi-carrier modulation technique that effectively partitionsthe overall system bandwidth into multiple (N_(F)) orthogonal subbands.These subbands are also referred to as tones, subcarriers, bins, andfrequency channels. With OFDM, each subband is associated with arespective subcarrier that may be modulated with data. Up to N_(F)modulation symbols may be transmitted on the N_(F) subbands in each OFDMsymbol period. Prior to transmission, these modulation symbols aretransformed to the time-domain using an N_(F)-point inverse fast Fouriertransform (IFFT) to obtain a “transformed” symbol that contains N_(F)chips.

OFDM can be used to combat frequency selective fading, which ischaracterized by different channel gains at different frequencies of theoverall system bandwidth. It is well known that frequency selectivefading causes intersymbol interference (ISI), which is a phenomenonwhereby each symbol in a received signal acts as distortion to one ormore subsequent symbols in the received signal. The ISI distortiondegrades performance by impacting the ability to correctly detect thereceived symbols. Frequency selective fading can be convenientlycombated with OFDM by repeating a portion of each transformed symbol toform a corresponding OFDM symbol. The repeated portion is commonlyreferred to as a cyclic prefix.

The length of the cyclic prefix (i.e., the amount to repeat for eachOFDM symbol) is dependent on delay spread. The delay spread of awireless channel is the time span or duration of an impulse response forthe wireless channel. This delay spread is also the difference betweenthe earliest and latest arriving signal instances (or multipaths) at areceiver for a signal transmitted via the wireless channel by atransmitter. The delay spread of an OFDM system is the maximum expecteddelay spread of the wireless channels for all transmitters and receiversin the system. To allow all receivers in the system to combat ISI, thecyclic prefix length should be equal to or longer than the maximumexpected delay spread. However, since the cyclic prefix represents anoverhead for each OFDM symbol, it is desirable to have the cyclic prefixlength be as short as possible to minimize overhead. As a compromise,the cyclic prefix length is typically selected such that the cyclicprefix contains a significant portion of all multipath energies for mostreceivers in the system.

An OFDM system can withstand a delay spread that is smaller than orequal to the cyclic prefix length. When this is the case, the N_(F)subbands are orthogonal to one another. However, a given receiver in thesystem may observe excess delay spread, which is a delay spread that isgreater than the cyclic prefix length. Excess delay spread can causevarious deleterious effects, such as ISI and channel estimation errors,both of which can degrade system performance as described below. Thereis therefore a need in the art for techniques to mitigate thedeleterious effects of excess delay spread in an OFDM system.

SUMMARY

Techniques for transmitting pilot and estimating the response of awireless channel with excess delay spread are described herein. Tomitigate the deleterious effects of excess delay spread, the number ofpilot subbands is selected to be greater than the cyclic prefix length(i.e., N_(Peff)>N_(cp)) to achieve “oversampling” in the frequencydomain. The oversampling may be obtained by either (1) using more pilotsubbands in each OFDM symbol period or (2) using different sets of pilotsubbands in different OFDM symbol periods (i.e., staggered pilotsubbands). For example, a staggered pilot transmission scheme may usetwo sets of pilot subbands, with each set containing N_(cp) pilotsubbands. The pilot subbands in the first set are staggered or offsetfrom the pilot subbands in the second set.

In one exemplary channel estimation technique for the above staggeredpilot transmission scheme, a first group of received pilot symbols forthe first pilot subband set is obtained in a first symbol period andused to derive a first (initial) frequency response estimate for awireless channel. A second group of received pilot symbols for thesecond pilot subband set is obtained in a second symbol period and usedto derive a second (initial) frequency response estimate for thewireless channel. First and second channel impulse response estimatesare derived based on the first and second frequency response estimates,respectively. A third (full) channel impulse response estimate is thenderived based on (e.g., by repeating and either combining or filtering)the first and second channel impulse response estimates, as describedbelow. The third channel impulse response estimate contains more tapsthan the number of pilot subbands in either the first or second set,which permits a more accurate characterization of the wireless channelin the presence of excess delay spread. A third (final) frequencyresponse estimate is derived based on the third channel impulse responseestimate and may be used for detection and other purposes. The channelestimation may be tailored to the specific staggered pilot transmissionscheme selected for use.

Various aspects and embodiments of the invention are described infurther detail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The features and nature of the present invention will become moreapparent from the detailed description set forth below when taken inconjunction with the drawings in which like reference charactersidentify correspondingly throughout and wherein:

FIG. 1 shows an OFDM modulator for an OFDM system;

FIGS. 2A and 2D show a wireless channel with excess delay spread and itseffective channel, respectively;

FIGS. 2B and 2C show a sequence of received chips for the wirelesschannel;

FIG. 3 shows a subband structure that may be used for the OFDM system;

FIGS. 4A, 4B and 4C show a sampled channel for a wireless channel, itseffective channel, and its estimated channel with critical sampling,respectively;

FIGS. 5, 9A and 9B show three staggered pilot transmission schemes;

FIG. 6 shows a process for deriving a full channel impulse responseestimate based on the staggered pilot transmission scheme shown in FIG.5;

FIG. 7 shows the derivation of the full channel impulse responseestimate;

FIG. 8A shows an estimated channel with oversampling and truncation;

FIG. 8B shows an estimated channel with oversampling and no truncation;

FIG. 10 shows a process for performing channel estimation for a givenstaggered pilot transmission scheme;

FIG. 11 shows an access point and a terminal in the OFDM system; and

FIG. 12 shows a channel estimator.

DETAILED DESCRIPTION

The word “exemplary” is used herein to mean “serving as an example,instance, or illustration.” Any embodiment or design described herein as“exemplary” is not necessarily to be construed as preferred oradvantageous over other embodiments or designs.

FIG. 1 shows a block diagram of an OFDM modulator 100 for an OFDMsystem. The data to be transmitted is typically encoded and interleavedto generate code bits, which are then mapped to modulation symbols. Thesymbol mapping is performed by (1) grouping the code bits into B-bitbinary values, where B≧1, and (2) mapping each B-bit value to a specificmodulation symbol based on a modulation scheme (e.g., M-PSK or M-QAM,where M=2^(B)). Each modulation symbol is a complex value in a signalconstellation corresponding to the modulation scheme. For each OFDMsymbol period, one “transmit” symbol is sent on each of the N_(F)subbands. Each transmit symbol can be either a modulation symbol forpilot/data or a signal value of zero (i.e., a “zero symbol”). An IFFTunit 110 performs an N_(F)-point IFFT on the N_(F) transmit symbols forthe N_(F) total subbands in each OFDM symbol period and provides atransformed symbol that contains N_(F) chips. The IFFT may be expressedas:s=W _(N) _(F) _(×N) _(F) ^(H) S,  Eq (1)where

-   -   S is an N_(F)×1 vector of transmit symbols for the N_(F)        subbands;    -   W _(N) _(F) _(×N) _(F) is an N_(F)×N_(F) discrete Fourier        transform (DFT) matrix;    -   s is an N_(F)×1 vector of time-domain chips; and    -   “^(H)” denotes the conjugate transpose.        The DFT matrix W _(N) _(F) _(×N) _(F) is defined such that the        (n,m)-th entry, w_(n,m), is given as:

$\begin{matrix}{{w_{n,m} = {\mathbb{e}}^{{- {j2\pi}}\frac{{({n - 1})}{({m - 1})}}{N_{F}}}},\mspace{14mu}{{{for}\mspace{14mu} n} = {{\left\{ {1\mspace{14mu}\ldots\mspace{14mu} N_{F}} \right\}\mspace{14mu}{and}\mspace{14mu} m} = \left\{ {1\mspace{14mu}\ldots\mspace{14mu} N_{F}} \right\}}},} & {{Eq}\mspace{14mu}(2)}\end{matrix}$where n is a row index and m is a column index. W _(N) _(F) _(×N) _(F)^(H) is an inverse DFT matrix.

A cyclic prefix generator 120 repeats a portion of each transformedsymbol to obtain a corresponding OFDM symbol that contains N_(C) chips,where N_(C)=N_(F)+N_(cp) and N_(cp) is the cyclic prefix length. An OFDMsymbol period is the duration of one OFDM symbol, which is N_(C) chipperiods. The chips are conditioned and transmitted via a wirelesschannel.

FIG. 2A shows an exemplary impulse response 210 of a wireless channelwith excess delay spread. Channel impulse response 210 includes two taps212 and 214 for two multipaths in the wireless channel. Tap 212 has acomplex gain of h₁ and is located at tap index 1. Tap 214 has a complexgain of h_(e) and is located at tap index N_(e), which is outside of thecyclic prefix length N_(cp). As used herein, “main channel” refers tothe portion of the channel impulse response that is at or within thecyclic prefix length, “excess channel” refers to the portion of thechannel impulse response that is outside of the cyclic prefix length,and “excess” refers to the difference between the tap index of an excesschannel tap and the cyclic prefix length. For channel impulse response210, the main channel includes one tap 212, the excess channel includesone tap 214, and the excess for tap 214 is N_(ex)=N_(e)−N_(cp).

FIG. 2B shows a sequence 220 of received chips for the wireless channelshown in FIG. 2A. Received chip sequence 220 is a convolution of atransmitted chip sequence with taps 212 and 214 for the wirelesschannel. Received chip sequence 220 is composed of (1) a chip sequence222 generated by convolving main channel tap 212 with the transmittedchip sequence and (2) a chip sequence 224 generated by convolving excesschannel tap 214 with the transmitted chip sequence, where s_(i) denotesthe i-th chip for the current OFDM symbol, x_(i) denotes the i-th chipfor the previous OFDM symbol, and i=1 . . . N_(C).

FIG. 2C shows the decomposition of received chip sequence 220 intodifferent components. Chip sequence 224 in FIG. 2B is replaced with (1)a chip sequence 226 generated by a circular convolution of excesschannel tap 214 with the N_(C) chips for the current OFDM symbol, (2) achip sequence 228 for the tail end of the previous OFDM symbol, and (3)a chip sequence 230 for the tail end of the current OFDM symbol. Chipsequences 222 and 226 represent the sequences that would have beenreceived for taps 212 and 214 if the cyclic prefix length weresufficiently long and tap 214 is part of the main channel. However,since this is not the case, chip sequences 228 and 230 are both due tothe excess delay spread. Chip sequence 228 represents the leakage of theprevious OFDM symbol into the current OFDM symbol and is the source ofintersymbol interference. Chip sequence 230 represents the disturbanceto the circular convolution and is the source of intercarrierinterference (ICI) and channel attenuation.

The intersymbol interference observed in each subband may be expressedas:ISI(k)=h _(e) ·W _(1×N) _(ex) (k) W _(N) _(ex) _(×N) _(F) ^(H) X,for k=1. . . N _(F),  Eq (3)where

-   -   X is an N_(F)×1 vector of transmit symbols for the previous OFDM        symbol;    -   W _(N) _(ex) _(×N) _(F) ^(H) is an N_(ex)×N_(F) matrix with the        last N_(ex) rows of W_(N) _(F) _(×N) _(F) ^(H); and    -   W _(1×N) _(ex) (k) is a 1×N_(ex) vector with the first N_(ex)        elements of the k-th row of W _(N) _(F) _(×N) _(F) .        The operation W _(N) _(ex) _(×N) _(F) ^(H) X generates an        N_(ex)×1 vector X _(N) _(ex) that contains the last N_(ex) chips        of the previous OFDM symbol. The multiplication of X _(N) _(ex)        with W _(1×N) _(ex) (k) generates the interference due to these        last N_(ex) chips on subband k.

The noise power on each subband due to intersymbol interference can beexpressed as:σ_(ISI) ² =E _(S) ·|h _(e)|²·(N _(ex) /N _(F)),for k=1 . . . N _(F),  Eq(4)where E_(S) is the transmit symbol energy, |h_(e)|² is the power of theexcess channel, and σ_(ISI) ² is the noise power due to ISI on eachsubband. As shown in equation (4), the ISI noise power per subband is(1) proportional to the excess channel energy |h_(e)|², (2) proportionalto the excess N_(ex), which is indicative of the amount of leakage ofthe previous OFDM symbol onto the current OFDM symbol, and (3) inverselyrelated to the number of total subbands since the total ISI noise poweris distributed over the N_(F) subbands.

The noise power on each subband due to intercarrier interference can becomputed in similar manner as for intersymbol interference and expressedas:σ_(ICI) ² =E _(S) ·|h _(e)|²·[(N _(ex) /N _(F))−(N _(ex) /N _(F))²],fork=1 . . . N _(F),  Eq (5)where σ_(ICI) ² is the noise power due to ICI on each subband.

FIG. 2D shows an “effective” channel 240 for the wireless channel shownin FIG. 2A. Referring back to FIG. 2C, chip sequence 226 represents thecontribution due to excess channel tap 214 (assuming that the cyclicprefix is long enough), and chip sequence 230 represents the source ofICI due to the excess channel. The subtraction operation for chipsequence 230 results partly in a reduction of the signal power for eachsubband. This subtraction can be accounted for by scaling down excesschannel tap 214 by a factor of (1−N_(ex)/N_(F)). As shown in FIG. 2D,effective channel 240 includes tap 212 having the complex gain of h₁ anda tap 216 having a complex gain of h_(e)·(1−N_(ex)/N_(F)). The reductionin the gain of tap 216 relative to the gain of tap 214 is referred to as“channel attenuation” and results from excess delay spread for tap 214.The amount of attenuation is related to the excess N_(ex).

A receiver performs channel estimation in order to derive a channelestimate for the wireless channel. Channel estimation is typicallyperformed based on pilot symbols, which are modulation symbols that areknown a priori by the receiver. The pilot symbols may be transmitted invarious manners as described below.

FIG. 3 shows an exemplary subband structure that may be used for theOFDM system. The OFDM system has an overall system bandwidth of BW MHz,which is partitioned into N_(F) orthogonal subbands using OFDM. Eachsubband has a bandwidth of BW/N_(F) MHz. For a spectrally shaped OFDMsystem, only N_(U) of the N_(F) total subbands are used for data/pilottransmission, where N_(U)<N_(F), and the remaining N_(F)−N_(U) subbandsare not used for data/pilot transmission and serve as guard subbands toallow the system to meet spectral mask requirements. For simplicity, thefollowing description assumes that all N_(F) subbands may be used in theOFDM system.

FIG. 3 also shows an exemplary frequency division multiplex (FDM) pilottransmission scheme 300. N_(P) subbands are used for pilot transmissionand are referred to as “pilot subbands”. To simplify computation for thechannel estimate, N_(P) may be selected as a power of two, and the N_(P)pilot subbands may be uniformly distributed across the N_(F) totalsubbands such that consecutive pilot subbands are spaced apart byN_(F)/N_(P) subbands.

The receiver can derive an initial frequency response estimate of thewireless channel based on received pilot symbols for the pilot subbands,as follows:

$\begin{matrix}{{{{\hat{H}}_{p}(k)} = \frac{y_{p}(k)}{p(k)}},\mspace{14mu}{{{for}\mspace{14mu} k} \in K_{p}},} & {{Eq}\mspace{14mu}(6)}\end{matrix}$where

-   -   y_(p)(k) is a received pilot symbol for subband k;    -   p(k) is a pilot symbol transmitted on subband k;    -   Ĥ_(p)(k) is a channel gain estimate for pilot subband k; and    -   K_(p) is a set of pilot subbands.        An N_(P)×1 vector Ĥ _(p) for the initial frequency response        estimate for N_(P) uniformly distributed pilot subbands may be        formed as Ĥ _(p)=[Ĥ_(p)(1) Ĥ_(p)(2) . . . Ĥ_(p)(N_(P))]^(T),        where “^(T)” denotes the transpose. If pilot symbols are not        transmitted on any one of the N_(P) pilot subbands (e.g., for a        spectrally shaped OFDM system), then extrapolation and/or        interpolation may be performed as necessary to obtain channel        gain estimates for pilot subbands without pilot transmission.        Filtering may also be performed on the vectors Ĥ _(p), obtained        for different OFDM symbol periods to improve the quality of the        initial frequency response estimate.

The frequency response estimate for the N_(F) total subbands may beobtained based on the initial frequency response estimate Ĥ _(p) usingvarious techniques. For a least-squares channel estimation technique, aleast-squares impulse response estimate for the wireless channel isfirst obtained as follows:ĥ _(N) _(P) =W _(N) _(P) _(×N) _(P) ^(H) Ĥ _(p),  Eq (7)where

-   -   W _(N) _(P) _(×N) _(P) is an N_(P)×N_(P) DFT matrix for the        N_(P) pilot subbands; and    -   ĥ _(N) _(P) is an N_(P)×1 vector for the least-squares impulse        response estimate.        Equation (7) indicates that the maximum number of channel taps        that can be estimated is limited to the number of pilot subbands        (i.e., N_(tap)=N_(P)).

The vector ĥ _(N) _(P) can be post-processed, for example, by settingtaps with values less than a predetermined threshold to zero, settingtaps for the excess channel to zero, and so on, as described below. Thevector ĥ _(N) _(P) is then zero-padded to length N_(F). The zero-paddedvector ĥ _(N) _(F) is transformed with an N_(F)-point FFT to obtain avector Ĥ _(N) _(F) for the final frequency response estimate, asfollows:Ĥ _(N) _(F) =W _(N) _(F) _(×N) _(F) ĥ _(N) _(F) ,  Eq (8)where

-   -   Ĥ _(N) _(F) =[Ĥ(1) Ĥ(2) . . . Ĥ(N_(F))]^(T).

FIG. 4A shows a generic impulse response 410 for a wireless channel.Channel impulse response 410 includes (1) N_(cp) taps with indices of 1through N_(cp) for the main channel and (2) L taps with indices ofN_(cp)+1 through N_(cp)+L for the excess channel. L is the time span orlength of the excess channel and is greater than zero when excess delayspread is present. Each tap has a complex gain of h_(i), which ingeneral may be a non-zero or zero value.

FIG. 4B shows an impulse response 420 for an effective channel for thewireless channel in FIG. 4A. Channel impulse response 420 includes allof the taps of channel impulse response 410. However, each of the L tapsfor the excess channel is scaled by a scaling factor of α_(N) _(i)=(1−N_(i)/N_(F)), where N_(i) is the excess for the tap and N_(i)=1 . .. L. The time span of the effective channel is equal to the time span ofthe wireless channel and is greater than the cyclic prefix length in thepresence of excess delay spread. The frequency response for the wirelesschannel can be obtained by performing an FFT on impulse response 420 forthe effective channel.

The channel impulse response for the effective channel can be estimatedbased on the received pilot symbols, as shown in equations (6) and (7).The accuracy of the channel impulse response estimate is impacted by thenumber of pilot subbands.

For a critically-sampled OFDM system, the number of pilot subbands isequal to the cyclic prefix length (i.e., N_(P)=N_(cp)). Since the numberof pilot subbands determines the maximum time span that can be estimatedfor the channel impulse response, up to N_(cp) channel taps for indicesof 1 through N_(cp) can be estimated for the critically-sampled system.

FIG. 4C shows an impulse response 430 for an estimated channel for thecritically-sampled OFDM system with excess delay spread. The time spanof the effective channel is longer than the cyclic prefix length whenexcess delay spread is present. In this case, the excess channel taps atindices of N_(cp)+1 through N_(cp)+L cannot be estimated because aninsufficient number of degrees of freedom exists for thecritically-sampled OFDM system. Furthermore, the channel impulseresponse for the wireless channel is undersampled in the frequencydomain by the N_(P) pilot subbands. This then causes a wrap aroundeffect of the excess channel in the time domain so that the excesschannel tap at index N_(cp)+1 appears at index 1, the excess channel tapat index N_(cp)+2 appears at index 2, and so on. Each wrap around excesschannel tap causes an error in estimating the corresponding main channeltap.

If an FFT is performed on channel impulse response 430, then theresultant frequency response estimate for each subband can be expressedas:Ĥ _(cs)(k)=H(k)+H _(err)(k),for k=1 . . . N _(F),  Eq (9)where

-   -   H(k) is the actual channel gain for subband k;    -   Ĥ_(cs)(k) is the channel gain estimate for subband k with        critical sampling; and    -   H_(err)(k) is the error in the channel gain estimate for subband        k.        For simplicity, channel gain error due to other noise is not        shown in equation (9).

The channel gain error H_(err)(k) can be expressed as:

$\begin{matrix}{{{H_{err}(k)} = {2\;{{\mathbb{e}}^{{j\pi}{({\frac{N_{cp}k}{N_{F}} + \frac{1}{2}})}} \cdot {\sin\left( \frac{\pi \cdot N_{cp} \cdot k}{N_{F}} \right)} \cdot {H_{ex}(k)}}}},\mspace{14mu}{{{for}\mspace{14mu} k} = {1\mspace{14mu}\ldots\mspace{14mu} N_{F}}},} & {{Eq}\mspace{14mu}(10)}\end{matrix}$where H_(ex)(k) is the complex gain for subband k due to the excesschannel, which can be obtained by performing an FFT on the excesschannel taps. The channel gain error H_(err)(k) can be decomposed intofour parts. The factor of 2 immediately to the right of the equal signin equation (10) reflects the two sources of channel gain error: (1) theinability to sample the excess channel and (2) the wrap around of theexcess channel onto the main channel The sine term corresponds to asinusoidal having a frequency determined by the ratio of N_(cp) overN_(F). The total noise power for the channel gain errors for allsubbands may be expressed as:

$\begin{matrix}{\begin{matrix}{{\sigma_{ch}^{2}(k)} = {\sum\limits_{k = 1}^{N_{F}}\;{{H_{err}(k)}}^{2}}} \\{{{= {2 \cdot {\sum\limits_{k = 1}^{N_{F}}{{{H_{ex}(k)}}^{2} \cdot \left( {1 - {\cos\left( \frac{\pi \cdot N_{cp} \cdot k}{N_{F}} \right)}} \right)}}}},}\mspace{14mu}}\end{matrix}{{{for}\mspace{14mu} k} = {1\mspace{14mu}\ldots\mspace{14mu}{N_{F}.}}}} & {{Eq}\mspace{14mu}(11)}\end{matrix}$

The signal-to-noise-and-interference ratio (SNR) for each subband may beexpressed as:

$\begin{matrix}{{{{SNR}(k)} = \frac{E_{S} \cdot {\underset{\_}{h}}^{2}}{N_{0} + {E_{S} \cdot \left\lbrack {{\sigma_{ch}^{2}(k)} + {\sigma_{ISI}^{2}(k)} + {\sigma_{ICI}^{2}(k)}} \right\rbrack}}},} & {{Eq}\mspace{14mu}(12)}\end{matrix}$where N₀ is the channel noise (which includes thermal noise,interference from other sources, receiver noise, and so on) and ∥h∥² isthe 2-norm of the effective channel impulse response. As shown inequation (12), the channel estimation error, ISI, and ICI noise powersare all scaled by the signal power E_(S). These three noise terms thusmanifest as a noise floor for the SNR. The noise floor due to channelestimation error, ISI, and ICI noise powers may be neglected if they arelower than the channel noise N₀. However, this noise floor may limit theperformance of the system if these noise powers are higher than thechannel noise N₀. The channel estimation error noise power may dominatethe ISI and ICI noise powers if the excess channel taps contain asignificant portion (e.g., 10% or more) of the total channel energy.

To mitigate the deleterious effects of excess delay spread on channelestimation error and SNR, the number of pilot subbands may be increased.For an over-sampled OFDM system, the “effective” number of pilotsubbands (which is the number of different pilot subbands used forchannel estimation) is greater than the cyclic prefix length (i.e.,N_(Peff)>N_(cp)). If N_(Peff) is sufficiently large so that the impulseresponse of the wireless channel (including the excess channel) does notexceed N_(Peff) taps, then a sufficient number of degrees of freedom isavailable to estimate all of the taps for the wireless channel in thepresence of excess delay spread.

Additional pilot subbands for oversampling may be obtained by variousmeans. In one pilot transmission scheme, N_(Peff)=N_(P)>N_(cp) and pilotsymbols are transmitted on all N_(P) pilot subbands in each OFDM symbolperiod. To simplify computation, N_(P) may be selected to be a power oftwo (e.g., N_(P)=2N_(cp)) and the N_(P) pilot subbands may be uniformlydistributed across the N_(F) total subbands. Fewer subbands would beavailable for data transmission for this pilot transmission scheme.

FIG. 5 shows a staggered pilot transmission scheme 500 that may be usedto increase the effective number of pilot subbands without increasingpilot overhead. For scheme 500, N_(P)=N_(cp) pilot subbands are used foreach OFDM symbol period. However, the N_(cp) pilot subbands for odd OFDMsymbol periods are staggered or offset from the N_(cp) pilot subbandsfor even OFDM symbol periods by N_(F)/2N_(cp) subbands. Scheme 500 usestwo different sets of N_(cp) pilot subbands, which corresponds to arepetition factor of two. The effective number of pilot subbands is thusN_(Peff)=2N_(P)=2N_(cp). To simplify computation, the N_(cp) pilotsubbands for each OFDM symbol may be uniformly distributed across theN_(F) total subbands.

FIG. 6 shows a process 600 for deriving a full channel impulse responseestimate of length N_(Peff)=2N_(cp) for a wireless channel based onpilot transmission scheme 500. An initial frequency response estimate Ĥ_(p0) is obtained based on received pilot symbols for the first set ofN_(cp) pilot subbands used in OFDM symbol period n, as shown in equation(6) (block 612). An initial frequency response estimate Ĥ _(p1) is alsoobtained based on received pilot symbols for the second set of N_(cp)pilot subbands used in OFDM symbol period n+1 (block 614). AnN_(cp)-point IFFT is performed on Ĥ _(p0) to obtain a channel impulseresponse estimate ĥ ₀ with N_(cp) taps (block 616). An N_(cp)-point IFFTis also performed on Ĥ _(p1) to obtain another channel impulse responseestimate ĥ ₁ with N_(cp) taps (block 618). For scheme 500 with arepetition of two, the vector ĥ ₀ is repeated to obtain a vector ĥ′₀ oflength N_(Peff)=2N_(cp) (block 620). The vector ĥ ₁ is also repeated butfurther phase adjusted to obtain a vector ĥ′₁ of length N_(Peff) (alsoblock 620). The vectors ĥ′₀ and ĥ′₁ are then combined (e.g., filtered)to obtain a full channel impulse response estimate ĥ _(N) _(Peff) withN_(Peff) taps (block 622). The vector ĥ _(N) _(Peff) may be furtherprocessed (e.g., to suppress noise) and is zero-filled to obtain avector ĥ _(N) _(F) of length N_(F) (block 624). An N_(F)-point FFT isthen performed on the vector ĥ _(N) _(F) to obtain the final frequencyresponse estimate Ĥ _(N) _(F) for the N_(F) subbands, as shown inequation (8) (block 626).

FIG. 6 shows an embodiment whereby the channel estimates for the twosets of pilot subbands are combined in the time domain. This is achievedby (1) deriving an initial channel impulse response estimate for theinitial frequency response estimate for each set of pilot subbands(blocks 616 and 618) and (2) combining the initial channel impulseresponse estimates for the two sets of pilot subbands to obtain the fullchannel impulse response estimate (block 622). The initial frequencychannel response estimates for the two sets of pilot subbands may alsobe combined in the frequency domain to obtain an intermediate frequencyresponse estimate, which may then be used to derive the full channelimpulse response estimate.

FIG. 7 illustrates the derivation of the full channel impulse responseestimate ĥ _(N) _(Peff) with N_(Peff)=2N_(cp) taps based on staggeredpilot transmission scheme 500. The vector ĥ ₀ represents a channelimpulse response estimate with N_(cp) taps and includes (1) a response712 for the main channel and (2) a response 714 for the wrap aroundexcess channel, which is caused by undersampling in the frequency domainwith N_(cp) pilot subbands. The vector ĥ ₀ is repeated to obtain avector ĥ′₀=[ĥ ₀ ĥ ₀]^(T). The vector ĥ ₁ similarly includes a response722 for the main channel and a response 724 for the wrap around excesschannel. The vector ĥ ₁ is also repeated, with the repeated instancebeing inverted, to obtain a vector ĥ′₁=[ĥ ₁−ĥ ₁]^(T). The vector ĥ _(N)_(Peff) may be obtained by summing the vectors ĥ′₀ and ĥ′₁, as shown inFIG. 7. The vector ĥ _(N) _(Peff) may also be obtained by filtering thevectors ĥ′₀ and ĥ′₁, as described below.

The vector ĥ _(N) _(Peff) represents the full channel impulse responseestimate with N_(Peff)=2·N_(cp) taps and includes (1) a response 732 forthe main channel, (2) a response 734 for the uncanceled portion of thewrap around excess channel, (3) a response 736 for the excess channel,and (4) a response 738 for the uncanceled portion of the main channel.Responses 734 and 738 may be due to various factors such as, forexample, changes in the wireless channel between the times that thevectors ĥ ₀ and ĥ ₁ are obtained.

As shown in FIG. 7, the full channel impulse response (with N_(Peff)taps) of the wireless channel can be estimated based on two receivedOFDM symbols each containing N_(cp) pilot subbands. If the wirelesschannel is relatively static over the two OFDM symbols, then responses734 and 738 may be small and the vector ĥ _(N) _(Peff) is an accuratefull impulse response estimate of the wireless channel.

The full channel impulse response estimate ĥ _(N) _(Peff) may be used invarious manners to obtain the final frequency response estimate Ĥ _(N)_(F) . All or some of the taps in ĥ _(N) _(Peff) may be selected foruse, and zero or more of the taps may be set to zero (i.e., zeroed out)to suppress noise. Several tap selection schemes are described below.

FIG. 8A shows an impulse response 810 for an estimated channel for afirst tap selection scheme. For this scheme, the first N_(cp) taps (forthe main channel) of the full channel impulse response estimate ĥ _(N)_(Peff) are used and the last N_(Peff)−N_(cp) taps (for the excesschannel) are set to zero (i.e., truncated). Estimated channel impulseresponse 810 thus suffers a truncation effect since the excess channelresponse has been zeroed out. However, impulse response 810 does notexperience wrap around effect. The channel estimation error for this tapselection scheme is determined by the excess channel and may beexpressed as:H _(err,tr)(k)=H _(ex)(k),for k=1 . . . N _(F).  Eq (13)

The channel estimation error noise power for this scheme is on the orderof the excess channel energy and is approximately half of the noisepower for the critically-sampled case shown in equation (11). For thefirst tap selection scheme, the truncation effect presents a noise floorfor SNR but the wrap around effect is not present and does not affectthe noise floor. Thus, the noise floor for the first tap selectionscheme is lower than that for the critically-sampled case.

The first tap selection scheme also provides an “oversampling gain”,which is a reduction in noise resulting from zeroing out some of thetaps. Since the last N_(Peff)−N_(cp) taps are set to zero, they do notintroduce any noise and do not degrade the final frequency responseestimate Ĥ _(N) _(F) . If N_(Peff)=2N_(cp) and the last N_(cp) taps arezeroed out, then the noise is reduced by approximately 3 dB over thecritically-sampled case.

FIG. 8B shows an impulse response 820 for an estimated channel for asecond tap selection scheme. For this scheme, all N_(Peff) taps for thefull channel impulse response estimate ĥ _(N) _(Peff) are used.Estimated channel impulse response 820 does not experience truncationeffect or wrap around effect since the excess channel response isproperly estimated with a sufficient number of pilot subbands. As aresult, the channel estimation error noise power for this scheme isapproximately zero and the SNR does not observe a noise floor due tothese two effects. However, since all N_(Peff) taps are used, noreduction in noise (i.e., no oversampling gain) is achieved over thecritically-sampled case.

Table 1 summarizes the effects observed for the critical sampling andoversampling cases. A ‘yes’ in the Truncate column indicates that thelast N_(Peff)−N_(cp) taps of the full channel impulse response estimateĥ _(N) _(Peff) are set to zero, and a ‘no’ indicates that all N_(Peff)taps are used.

TABLE 1 Trun- Wrap Around Truncation Oversampling Sampling cate EffectEffect Gain Critical Sampling — Yes Yes No (N_(Peff) = N_(cp))Oversampling Yes No Yes Yes (N_(Peff) > N_(cp)) No No No No

The first and second tap selection schemes select taps in adeterministic manner. The tap selection may also be performed in othermanners, some of which are described below.

In a third tap selection scheme, “thresholding” is used to selectchannel taps with sufficient energy and to zero out channel taps withlow energy. Channel taps with low energy are likely due to noise ratherthan signal energy. A threshold may be used to determine whether or nota given channel tap has sufficient energy and should be retained. Thethreshold may be computed based on various factors and in variousmanners. The threshold may be a relative value (i.e., dependent on themeasured channel response) or an absolute value (i.e., not dependent onthe measured channel response). A relative threshold may be computedbased on the (e.g., total or average) energy of the channel impulseresponse estimate. The use of the relative threshold ensures that (1)the thresholding is not dependent on variations in the received energyand (2) the channel taps that are present but having low signal energyare not zeroed out. An absolute threshold may be computed based on thenoise at the receiver, the lowest energy expected for the received pilotsymbols, and so on. The use of the absolute threshold forces the channeltaps to meet some minimum value in order to be selected for use. Thethreshold may also be computed based on a combination of factors usedfor relative and absolute thresholds. For example, the threshold may becomputed based on the energy of the channel impulse response estimateand further constrained to be equal to or greater than a predeterminedminimum value.

The thresholding may be performed in various manners. In onethresholding scheme, the thresholding is performed after the truncationof the last N_(Peff)−N_(cp) taps and may be expressed as:

$\begin{matrix}{{\hat{h}}_{i} = \left\{ {{{\begin{matrix}0 & {{{{for}\mspace{14mu}{{\hat{h}}_{i}}^{2}} < E_{th}},} \\{\hat{h}}_{i} & {otherwise}\end{matrix}\mspace{14mu}{for}\mspace{14mu} i} = {1\mspace{14mu}\ldots\mspace{14mu} N_{cp}}},} \right.} & {{Eq}\mspace{14mu}(14)}\end{matrix}$where

-   -   ĥ_(i) is the i-th element/tap in ĥ _(N) _(Peff) ;    -   |ĥ_(i)|² is the energy of the i-th tap;    -   E_(th) is the threshold used to zero out low energy taps.        The threshold may be defined, for example, based on the energy        of the N_(cp) taps for the main channel as follows:        E_(th)=α_(th)·∥ĥ _(N) _(Peff) ∥², where ∥ĥ _(N) _(Peff) ∥² is        the main channel energy (after truncation) and α_(th) is a        coefficient. The coefficient α_(th) may be selected based on a        trade off between noise suppression and signal deletion. A        higher value for α_(th) provides more noise suppression but also        increases the likelihood of a low energy tap being zeroed out.        The coefficient α_(th) may be a value within a range of 0 to        1/N_(cp) (e.g., α_(th)=0.1/N_(cp)).

In another thresholding scheme, the thresholding is performed on allN_(Peff) elements of ĥ _(N) _(Peff) (i.e., without truncation) using asingle threshold, similar to that shown in equation (14). In yet anotherthresholding scheme, the thresholding is performed on all N_(Peff)elements of ĥ _(N) _(Peff) using multiple thresholds. For example, afirst threshold may be used for the first N_(cp) taps in ĥ _(N) _(Peff)for the main channel, and a second threshold may be used for the lastN_(Peff)−N_(cp) taps in ĥ _(N) _(Peff) for the excess channel. Thesecond threshold may be set lower than the first threshold. In yetanother thresholding scheme, the thresholding is performed on only thelast N_(Peff)−N_(cp) taps in ĥ _(N) _(Peff) and not on the first N_(cp)taps. The thresholding may be performed in other manners, and this iswithin the scope of the invention.

Thresholding is well suited for a wireless channel that is “sparse”,such as a wireless channel in a macro-cellular broadcast system. Asparse wireless channel has much of the channel energy concentrated in afew taps. Each tap corresponds to a resolvable signal path with adifferent propagation delay. A sparse channel includes few signal pathseven though the delay spread (i.e., time difference) between thesesignal paths may be large. The taps corresponding to weak ornon-existing signal paths can be zeroed out.

It can be shown that system performance may be improved significantly byoversampling with N_(Peff)>N_(cp). Oversampling in combination withtruncation of the last N_(Peff)−N_(cp) taps provides (1) a lower noisefloor in SNR because the wrap around effect is not present and (2) noisereduction due to oversampling gain. Oversampling without truncationremoves the noise floor due to wrap around and truncation effects butdoes not provide oversampling gain. Oversampling in combination withthresholding (with or without truncation) can provide furtherimprovement in certain scenarios. Truncation and/or thresholding mayalso be disabled or enabled based on the detected delay spread. Forexample, if the excess delay spread condition is detected (e.g., byperforming correlation on the received chips), then truncation may bedisabled and thresholding may be enabled or disabled. In any case,oversampling allows the receiver to obtain the full channel impulseresponse estimate, which can provide a more accurate channel estimateand improve system performance. In general, the amount of improvementwith oversampling increases as the amount of energy in the excesschannel increases.

FIG. 5 shows an exemplary staggered pilot transmission scheme with twosets of interlaced pilot subbands. Various other pilot transmissionschemes may also be used to obtain the necessary effective number ofpilot subbands for oversampling.

FIG. 9A shows a staggered pilot transmission scheme 910 with fourdifferent sets of pilot subbands. Each of the four sets includes N_(Psb)pilot subbands. To simplify computation, N_(Psb) may be selected to be apower of two, and the N_(Psb) pilot subbands in each set may beuniformly distributed across the N_(F) total subbands such thatconsecutive pilot subbands in each set are spaced apart by N_(F)/N_(Psb)subbands. For example, N_(Psb) may be equal to N_(cp), N_(cp)/2, and soon. The pilot subbands in the four sets are also interlaced in acomb-like structure, as shown in FIG. 9A. The four pilot subband setsare used in four OFDM symbol periods, for example, in the order shown inFIG. 9A or in a different order.

The received pilot symbols for the four sets of pilot subbands may beused in various manners for channel estimation. A channel impulseresponse estimate of length N_(Psb), 2N_(Psb), or 4N_(Psb) may beobtained based on the received pilot symbols for these four pilotsubband sets. A channel impulse response estimate of lengthN_(Peff)=2N_(Psb) may be obtained by (1) performing an N_(Psb)-pointIFFT on the N_(Psb) received pilot symbols for each OFDM symbol periodto obtain an impulse response estimate ĥ _(N) _(Psb) of length N_(Psb),(2) repeating the impulse response estimate ĥ _(N) _(Psb) once andadjusting the phase of each instance of ĥ _(N) _(Psb) as necessary toobtain a vector ĥ′_(2N) _(Psb) , and (3) updating the full channelimpulse response estimate ĥ _(N) _(Peff) with the vector ĥ′_(2N) _(Psb). A channel impulse response estimate of length N_(Peff)=4N_(Psb) may beobtained by (1) performing an N_(Psb)-point IFFT on the N_(Psb) receivedpilot symbols for each OFDM symbol period to obtain the impulse responseestimate ĥ _(N) _(Psb) , (2) repeating the impulse response estimate ĥ_(N) _(Psb) three times and adjusting the phases of each instance of ĥ_(N) _(Psb) as necessary to obtain a vector ĥ _(4N) _(Psb) , and (3)updating the full channel impulse response estimate ĥ _(N) _(Peff) withthe vector ĥ′_(4N) _(Psb) . The phase adjustment is dependent on thenumber of pilot subband sets and the pilot subbands in each set.

FIG. 9B shows a staggered pilot transmission scheme 920 with threedifferent sets of pilot subbands. The first set includes 2N_(Psb) pilotsubbands, and the second and third sets each include N_(Psb) pilotsubbands. To simplify computation, N_(Psb) may be selected to be a powerof two, and the N_(Psb) or 2N_(Psb) pilot subbands in each set may beuniformly distributed across the N_(F) total subbands. The pilotsubbands in the three sets are also interlaced in a comb-like structure,as shown in FIG. 9B. The three pilot subband sets may be used in threeOFDM symbol periods, for example, in the order shown in FIG. 9B or in adifferent order.

In general, a staggered pilot transmission scheme uses different sets ofpilot subbands for different OFDM symbol periods, and the effectivenumber of pilot subbands is equal to the number of different subbandsused for pilot transmission. Any number of pilot subband sets (orrepetitions) may be used. A higher repetition generally corresponds to ahigher effective number of pilot subbands and also a longer channelestimation delay. Furthermore, any number of pilot subbands may be usedfor each set, and the sets may include the same or different numbers ofsubbands. It may be advantageous to cycle through and transmit pilotsymbols on as many of the N_(F) total subbands as possible. However,only a small number of (e.g., N_(cp)) subbands are used in each OFDMsymbol period in order to reduce pilot overhead.

FIG. 10 shows a process 1000 for performing channel estimation for agiven staggered pilot transmission scheme. Initially, a group ofreceived pilot symbols is obtained for a set of pilot subbands used forpilot transmission in the current OFDM symbol period n (block 1012). Aninitial frequency response estimate Ĥ _(p)(n) is derived for these pilotsubbands based on the received pilot symbols (block 1014). An initialchannel impulse response estimate ĥ(n) is then derived based on (e.g.,by performing an IFFT on) the initial frequency response estimate Ĥ_(p)(n) (block 1016). The initial channel impulse response estimate ĥ(n)is repeated once or possibly more times (block 1018). Each instance ofĥ(n) is appropriately adjusted, for example, in phase based on theparticular pilot subbands used in the current OFDM symbol period n (alsoblock 1018). The output of block 1018 is an extended channel impulseresponse estimate ĥ′(n) with more taps than ĥ(n).

The full channel impulse response estimate ĥ _(N) _(Peff) (n) for thecurrent OFDM symbol period n is then updated based on ĥ′(n) (block1020). The updating of ĥ _(N) _(Peff) (n) may be performed in variousmanners depending on (1) the staggered pilot transmission schemeselected for use, (2) whether or not filtering is performed, and (3)possibly other factors. For example, if filtering is not performed andpilot transmission scheme 500 shown in FIG. 5 is used, then ĥ _(N)_(Peff) (n) may be set to ĥ′(n) for an odd-numbered OFDM symbol periodand computed as ĥ _(N) _(Peff) (n)=[ĥ _(N) _(Peff) (n−1)+ĥ′(n)]/2 for aneven-numbered OFDM symbol period. Filtering of ĥ′(n) to obtain ĥ _(N)_(Peff) (n) is described below. The full channel impulse responseestimate ĥ _(N) _(Peff) (n) may further be processed (e.g., truncated,threshold, and so on) and zero-filled to obtain a vector ĥ _(N) _(F) (n)of length N_(F) (block 1022). A final frequency response estimate Ĥ _(N)_(F) (n) for the current OFDM symbol period n is then derived based onthe channel impulse response estimate ĥ _(N) _(F) (n) (block 1024).Blocks 1012 through 1024 may be performed for each OFDM symbol period orwhenever pilot symbols are received.

As noted above, the full channel impulse response estimate ĥ _(N)_(Peff) (n) may be obtained by filtering ĥ′(n). For example, ĥ _(N)_(Peff) (n) may be obtained with a FIR filter as follows:

$\begin{matrix}{{{{\underset{\_}{\hat{h}}}_{N_{Peff}}(n)} = {\sum\limits_{i = {- L_{1}}}^{L_{2}}\;{{\underset{\_}{c}}_{i} \cdot {{\underset{\_}{\hat{h}}}^{\prime}\left( {n - i} \right)}}}},} & {{Eq}\mspace{14mu}(15)}\end{matrix}$where

-   -   c _(i) is a vector with N_(Peff) coefficients for FIR filter tap        i; and    -   L₁ and L₂ are the time extents of the FIR filter.        For a causal FIR filter, L₁=0, L₂≧1, and the filtered frequency        response estimate ĥ _(N) _(Peff) (n) is a weighted sum of the        extended channel impulse response estimates ĥ′(n) for L₂ prior        and the current OFDM symbol periods. For a non-causal FIR        filter, L₁≧1, L₂≧1, and the filtered frequency response estimate        ĥ _(N) _(Peff) (n) is a weighted sum of the extended channel        impulse response estimates ĥ′(n) for L₂ prior, the current, and        L₁ future OFDM symbol periods. Buffering of L₁ received OFDM        symbols is needed to implement the non-causal FIR filter.

The coefficients for the FIR filter may be selected in various manners.The L₁+L₂+1 vectors c _(i) for the L₁+L₂+1 taps of the FIR filter areselected to obtain the desired filtering characteristics (e.g., filterbandwidth and roll-off). The N_(Peff) coefficients for each vector c_(i) may also be selected in various manners. In one embodiment, theN_(Peff) coefficients in the vector c _(i) for each FIR filter tap areall set to the same value. In another embodiment, the first N_(cp)coefficients (for the main channel) in the vector c _(i) for each FIRfilter tap are set to one value, and the remaining N_(Peff)−N_(cp)coefficients are set to another value. In general, equal or differentweights may be used for the N_(Peff) coefficients in each vector c _(i).

The full channel impulse response estimate ĥ _(N) _(Peff) (n) may alsobe obtained with an IIR filter as follows:ĥ _(N) _(Peff) (n)=(1−α_(t))· ĥ _(N) _(Peff) (n−1)+α_(t) ·ĥ′(n),  Eq(16)where α_(t) is a time constant for the filtering. The time constantα_(t) may be selected based on the characteristics (e.g., coherencetime) of the wireless channel.

The initial frequency response estimate Ĥ _(p) (n) and/or the finalfrequency response estimate Ĥ _(N) _(F) (n) may also be filtered toobtain higher quality.

The final frequency response estimate Ĥ _(N) _(F) (n) may be used fordetection to recover the transmitted data symbols. The received symbolfor each subband may be expressed as:Y(k)=√{square root over (E _(S))}·{circumflex over (H)}(k)·S(k)+N(k),fork=1 . . . N _(F),  Eq (17)where

-   -   S(k) is the transmit symbol for subband k;    -   Ĥ(k) is the channel gain estimate for subband k;    -   N(k) is the noise observed for subband k; and    -   Y(k) is the received symbol for subband k.

The detection may be performed as follows:

$\begin{matrix}{{{\hat{S}(k)} = {\frac{Y(k)}{\hat{H}(k)} = {{S(k)} + {N^{\prime}(k)}}}},\mspace{14mu}{{{for}\mspace{14mu} k} \in K_{d}},} & {{Eq}\mspace{14mu}(18)}\end{matrix}$where

-   -   Ŝ(k) is a detected symbol on subband k;    -   N′(k) is the post-processed noise on subband k; and    -   K_(d) is a set of subbands used for data transmission (i.e., the        data subbands).        The operation in equation (18) is commonly referred to as        equalization and is typically used for an uncoded system.        Alternatively, the detection may be performed as:        Ŝ(k)=Y(k)Ĥ*(k)=S(k)+N″(k),for kεK _(d),  Eq (19)        where “*” denotes the complex conjugate. The operation in        equation (19) is commonly referred to as matched filtering and        is typically used for a coded system.

FIG. 11 shows a block diagram of an access point 1100 and a terminal1150 in the OFDM system. On the downlink, at access point 1100, atransmit (TX) data processor 1110 receives, formats, codes, interleaves,and modulates (i.e., symbol maps) traffic data and provides modulationsymbols (or simply, “data symbols”). An OFDM modulator 1120 receives thedata symbols and pilot symbols, performs OFDM modulation as describedfor FIG. 1, and provides a stream of OFDM symbols. Pilot symbols aretransmitted in a manner such that the effective number of pilot subbandsis greater than the cyclic prefix length (i.e., N_(Peff)>N_(cp)) toachieve oversampling. A transmitter unit (TMTR) 1122 receives andconverts the stream of OFDM symbols into one or more analog signals,conditions (e.g., amplifies, filters, and frequency upconverts) theanalog signals to generate a downlink signal, and transmits the signalvia an antenna 1124 to the terminals.

At terminal 1150, an antenna 1152 receives the downlink signal andprovides a received signal to a receiver unit (RCVR) 1154. Receiver unit1154 conditions (e.g., filters, amplifies, and frequency downconverts)the received signal, digitizes the conditioned signal, and providesreceived chips to an OFDM demodulator 1156.

FIG. 12 shows an embodiment of OFDM demodulator 1156. A cyclic prefixremoval unit 1212 removes the cyclic prefix appended to each OFDMsymbol. An FFT unit 1214 then transforms each received transformedsymbol to the frequency domain using an N_(F)-point FFT and obtainsN_(F) received symbols for the N_(F) subbands. FFT unit 1214 providesreceived pilot symbols to a processor 1170 and received data symbols toa detector 1216. Detector 1216 further receives a frequency responseestimate Ĥ _(N) _(F) _(,dn) for the downlink from processor 1170,performs detection on the received data symbols to obtain detectedsymbols (which are estimates of the transmitted data symbols), andprovides the detected symbols to an RX data processor 1158.

Processor 1170 includes a channel estimator 1220 that obtains thereceived pilot symbols and performs channel estimation as describedabove. Within channel estimator 1220, a pilot detector 1222 removes themodulation on the received pilot symbols and may perform extrapolationand/or interpolation as necessary to obtain an initial frequencyresponse estimate Ĥ _(p,dn) with channel gain estimates for N_(dn)uniformly distributed subbands in each OFDM symbol period. An IFFT unit1224 performs an IFFT on the initial frequency response estimate toobtain a channel impulse response estimate ĥ _(N) _(dn) _(,dn) withN_(dn) taps. A repetition unit 1226 repeats the channel impulse responseestimate as many times as necessary and further adjusts the phase ofeach instance if needed. A combiner/filter 1228 then either combines orfilters the output of unit 1226 and provides a full channel impulseresponse estimate. A threshold and zero-padding unit 1230 performsthresholding (if enabled) and zero-padding to obtain a vector ĥ _(N)_(F) _(,dn) with N_(F) taps. An FFT unit 1232 then performs an FFT onthe vector ĥ _(N) _(F) _(,dn) to obtain the final frequency responseestimate Ĥ _(N) _(F) _(,dn) for the N_(F) subbands for the downlink.

Referring back to FIG. 11, RX data processor 1158 demodulates (i.e.,symbol demaps), deinterleaves, and decodes the detected symbols torecover the transmitted traffic data. The processing by OFDM demodulator1156 and RX data processor 1158 is complementary to the processing byOFDM modulator 1120 and TX data processor 1110, respectively, at accesspoint 1100.

On the uplink, a TX data processor 1182 processes traffic data andprovides data symbols. An OFDM modulator 1184 receives and multiplexesthe data symbols with pilot symbols, performs OFDM modulation, andprovides a stream of OFDM symbols. The pilot symbols may be transmittedon N_(up) subbands that have been assigned to terminal 1150 for pilottransmission. The number of pilot subbands (N_(up)) for the uplink maybe the same or different from the number of pilot subbands (N_(dn)) forthe downlink. Moreover, the same or different (e.g., staggering) pilottransmission schemes may be used for the downlink and uplink. Atransmitter unit 1186 then receives and processes the stream of OFDMsymbols to generate an uplink signal, which is transmitted via anantenna 1152 to the access point.

At access point 1100, the uplink signal from terminal 1150 is receivedby antenna 1124 and processed by a receiver unit 1142 to obtain receivedchips. An OFDM demodulator 1144 then processes the received chips andprovides received pilot symbols and detected symbols for the uplink. AnRX data processor 1146 processes the detected symbols to recover thetraffic data transmitted by terminal 1150.

Processor 1130 performs channel estimation for each terminaltransmitting on the uplink, as described above. Multiple terminals maytransmit pilot concurrently on the uplink on their assigned pilotsubbands. To reduce interference, each subband may be used for pilot ordata transmission by only one terminal in a given OFDM symbol period.Processor 1130 may implement channel estimator 1220 shown in FIG. 12.For each terminal m, processor 1130 obtains an initial frequencyresponse estimate Ĥ _(m) for the uplink for the terminal based on pilotsymbols received from the terminal, derives a channel impulse responseestimate ĥ _(N) _(up) _(,m) for the terminal based on Ĥ _(m), andderives a final frequency response estimate Ĥ _(N) _(F) _(,m) for theterminal based on ĥ _(N) _(up) _(,m). The frequency response estimate Ĥ_(N) _(F) _(,m) for each terminal is provided to OFDM demodulator 1144and used for detection for that terminal.

Processors 1130 and 1170 direct the operation at access point 1100 andterminal 1150, respectively. Memory units 1132 and 1172 store programcodes and data used by processors 1130 and 1170, respectively.Processors 1130 and 1170 also perform channel estimation as describedabove.

For clarity, the pilot transmission and channel estimation techniqueshave been described for an OFDM system. These techniques may be used forother multi-carrier modulation techniques such as discrete multi tone(DMT).

The pilot transmission and channel estimation techniques describedherein may be implemented by various means. For example, thesetechniques may be implemented in hardware, software, or a combinationthereof. For a hardware implementation, the processing units used forchannel estimation may be implemented within one or more applicationspecific integrated circuits (ASICs), digital signal processors (DSPs),digital signal processing devices (DSPDs), programmable logic devices(PLDs), field programmable gate arrays (FPGAs), processors, controllers,micro-controllers, microprocessors, other electronic units designed toperform the functions described herein, or a combination thereof.

For a software implementation, the pilot transmission and channelestimation techniques may be implemented with modules (e.g., procedures,functions, and so on) that perform the functions described herein. Thesoftware codes may be stored in a memory unit (e.g., memory units 1132and 1172 in FIG. 11) and executed by a processor (e.g., processors 1130and 1170). The memory unit may be implemented within the processor orexternal to the processor, in which case it can be communicativelycoupled to the processor via various means as is known in the art.

The previous description of the disclosed embodiments is provided toenable any person skilled in the art to make or use the presentinvention. Various modifications to these embodiments will be readilyapparent to those skilled in the art, and the generic principles definedherein may be applied to other embodiments without departing from thespirit or scope of the invention. Thus, the present invention is notintended to be limited to the embodiments shown herein but is to beaccorded the widest scope consistent with the principles and novelfeatures disclosed herein.

1. A method for mitigating intersymbol interference and channelestimation errors in a wireless communication system, comprising:transmitting a first set of pilot subbands in a first symbol period; andtransmitting a second set of pilot subbands in a second symbol period,wherein the first set of pilot subbands and the second set of pilotsubbands are staggered in frequency, and the number of pilot subbandsselected are greater than a cyclic prefix length (N_(Peff)>N_(cp)) toachieve over sampling in the frequency domain.
 2. The method of claim 1,wherein the first set of pilot subbands and the second set of pilotsubbands include an equal number of pilot subbands.
 3. The method ofclaim 1, wherein the first set of pilot subbands and the second set ofpilot subbands include different numbers of pilot subbands.
 4. Themethod of claim 1, wherein the wireless communication system utilizesdiscrete multi tone (DMT).
 5. The method of claim 1, further comprising:transmitting a third set of pilot subbands in a third symbol period,wherein the third set of pilot subbands, the first set of pilot subbandsand the second set of pilot subbands are staggered in frequency.
 6. Anapparatus for mitigating intersymbol interference and channel estimationerrors in a wireless communication system, comprising: at least oneprocessor configured to transmit a first set of pilot subbands in afirst symbol period, and to transmit a second set of pilot subbands in asecond symbol period, wherein the first set of pilot subbands and thesecond set of pilot subbands are staggered in frequency, and wherein thenumber of pilot subbands selected are greater than a cyclic prefixlength (N_(Peff)>N_(cp)) to achieve oversampling in the frequencydomain; and a memory coupled to the at least one processor.
 7. Theapparatus of claim 6, wherein the first set of pilot subbands and thesecond set of pilot subbands include an equal number of pilot subbands.8. The apparatus of claim 6, wherein the first set of pilot subbands andthe second set of pilot subbands include different numbers of pilotsubbands.
 9. The apparatus of claim 6, wherein the wirelesscommunication system utilizes discrete multi tone (DMT).
 10. Theapparatus of claim 6, wherein the at least one processor is furtherconfigured to transmit a third set of pilot subbands in a third symbolperiod, wherein the third set of pilot subbands, the first set of pilotsubbands and the second set of pilot subbands are staggered infrequency.
 11. An apparatus for mitigating intersymbol interference andchannel estimation errors in a wireless communication system,comprising: means for transmitting a first set of pilot subbands in afirst symbol period; and means for transmitting a second set of pilotsubbands in a second symbol period, wherein the first set of pilotsubbands and the second set of pilot subbands are staggered infrequency, and the number of pilot subbands selected are greater than acyclic prefix length (N_(Peff)>N_(cp)) to achieve oversampling in thefrequency domain.
 12. The apparatus of claim 11, wherein the first setof pilot subbands and the second set of pilot subbands include an equalnumber of pilot subbands.
 13. The apparatus of claim 11, wherein thefirst set of pilot subbands and the second set of pilot subbands includedifferent numbers of pilot subbands.
 14. The apparatus of claim 11,wherein the wireless communication system utilizes discrete multi tone(DMT).
 15. The apparatus of claim 11, further comprising: means fortransmitting a third set of pilot subbands, in a third symbol period,wherein the third set of pilot subbands, the first set of pilot subbandsand the second set of pilot subbands are staggered in frequency.
 16. Acomputer program product, comprising a computer readable medium, formitigating intersymbol interference and channel estimation errors in awireless communication system, comprising: code for causing at least onecomputer to transmit a first set of pilot subbands in a first symbolperiod, and code for causing the at least one computer to transmit asecond set of pilot subbands in a second symbol period, wherein thefirst set of pilot subbands and the second set of pilots subbands arestaggered in frequency, and the number of pilot subbands selected aregreater than a cyclic prefix length (N_(Peff)>N_(cp)) to achieveoversampling in the frequency domain.
 17. The computer program productof claim 16, wherein the first set of pilot subbands and the second setof pilot subbands include an equal number of pilot subbands.
 18. Thecomputer program product of claim 16, wherein the first set of pilotsubbands and the second set of pilot subbands include different numbersof pilot subbands.
 19. The computer program product of claim 16, whereinthe wireless communication system utilizes discrete multi tone (DMT).20. The computer program product of claim 16, wherein thecomputer-readable medium further comprises: code for causing the atleast one computer to transmit a third set of pilot subbands in a thirdsymbol period, wherein the third set of pilot subbands, the first set ofpilot subbands and the second set of pilot subbands are staggered infrequency.
 21. A method for mitigating intersymbol interference andchannel estimation errors in a wireless communication system,comprising: receiving a first set of pilot subbands in a first symbolperiod; and receiving a second set of pilot subbands in a second symbolperiod, wherein the first set of pilot subbands and the second set ofpilot subbands are staggered in frequency, and the number of pilotsubbands selected are greater than a cyclic prefix length(N_(Peff)>N_(cp)) to achieve oversampling in the frequency domain. 22.The method of claim 21, wherein the first set of pilot subbands and thesecond set of pilot subbands include an equal number of pilot subbands.23. The method of claim 21, wherein the first set of pilot subbands andthe second set of pilot subbands include different numbers of pilotsubbands.
 24. The method of claim 21, wherein the wireless communicationsystem utilizes discrete multi tone (DMT).
 25. The method of claim 21,further comprising: receiving a third set of pilot subbands in a thirdsymbol period, wherein the third set of pilot subbands, the first set ofpilot subbands and the second set of pilot subbands are staggered infrequency.
 26. An apparatus for mitigating intersymbol interference andchannel estimation errors in a wireless communication system,comprising: at least one processor configured to receive a first set ofpilot subbands in a first symbol period, and tp receive a second set ofpilot subbands in a second symbol period, wherein the first set of pilotsubbands and the second set of pilot subbands are staggered infrequency, and wherein the number of pilot subbands selected are greaterthan a cyclic prefix length (N_(Peff)>N_(cp)) to achieve oversampling inthe frequency domain; and a memory coupled to the at least oneprocessor.
 27. The apparatus of claim 26, wherein the first set of pilotsubbands and the second set of pilot subbands include an equal number ofpilot subbands.
 28. The apparatus of claim 26, wherein the first set ofpilot subbands and the second set of pilot subbands include differentnumbers of pilot subbands.
 29. The apparatus of claim 26, wherein thewireless communication system utilizes discrete multi tone (DMT). 30.The apparatus of claim 26, wherein the at least one processor is furtherconfigured to receive a third set of pilot subbands in a third symbolperiod, wherein the third set of pilot subbands, the first set of pilotsubbands and the second set of pilot subbands are staggered infrequency.
 31. An apparatus for mitigating intersymbol interference andchannel estimation errors in a wireless communication system,comprising: means for receiving a first set of pilot subbands in a firstsymbol period; and means for receiving a second set of pilot subbands ina second symbol period, wherein the first set of pilot subbands and thesecond set of pilot subbands are staggered in frequency, and the numberof pilot subbands selected are greater than a cyclic prefix length(N_(Peff)>N_(cp)) to achieve oversampling in the frequency domain. 32.The apparatus of claim 31, wherein the first set of pilot subbands andthe second set of pilot subbands include an equal number of pilotsubbands.
 33. The apparatus of claim 31, wherein the first set of pilotsubbands and the second set of pilot subbands include different numbersof pilot subbands.